Non-volatile memory and method with compensation for source line bias errors

ABSTRACT

Source line bias is an error introduced by a non-zero resistance in the ground loop of the read/write circuits. During sensing the source of a memory cell is erroneously biased by a voltage drop across the resistance and results in errors in the applied control gate and drain voltages. This error is minimized when the applied control gate and drain voltages have their reference point located as close as possible to the sources of the memory cells. In one preferred embodiment, the reference point is located at a node where the source control signal is applied. When a memory array is organized in pages of memory cells that are sensed in parallel, with the sources in each page coupled to a page source line, the reference point is selected to be at the page source line of a selected page via a multiplexor.

FIELD OF THE INVENTION

This invention relates generally to non-volatile semiconductor memorysuch as electrically erasable programmable read-only memory (EEPROM) andflash EEPROM, and specifically ones having improved sensing circuitsthat compensate for source bias errors due to a finite resistance in theground loop.

BACKGROUND OF THE INVENTION

Solid-state memory capable of nonvolatile storage of charge,particularly in the form of EEPROM and flash EEPROM packaged as a smallform factor card, has recently become the storage of choice in a varietyof mobile and handheld devices, notably information appliances andconsumer electronics products. Unlike RAM (random access memory) that isalso solid-state memory, flash memory is non-volatile, and retaining itsstored data even after power is turned off. In spite of the higher cost,flash memory is increasingly being used in mass storage applications.Conventional mass storage, based on rotating magnetic medium such ashard drives and floppy disks, is unsuitable for the mobile and handheldenvironment. This is because disk drives tend to be bulky, are prone tomechanical failure and have high latency and high power requirements.These undesirable attributes make disk-based storage impractical in mostmobile and portable applications. On the other hand, flash memory, bothembedded and in the form of a removable card is ideally suited in themobile and handheld environment because of its small size, low powerconsumption, high speed and high reliability features.

EEPROM and electrically programmable read-only memory (EPROM) arenon-volatile memory that can be erased and have new data written or“programmed” into their memory cells. Both utilize a floating(unconnected) conductive gate, in a field effect transistor structure,positioned over a channel region in a semiconductor substrate, betweensource and drain regions. A control gate is then provided over thefloating gate. The threshold voltage characteristic of the transistor iscontrolled by the amount of charge that is retained on the floatinggate. That is, for a given level of charge on the floating gate, thereis a corresponding voltage (threshold) that must be applied to thecontrol gate before the transistor is turned “on” to permit conductionbetween its source and drain regions.

The floating gate can hold a range of charges and therefore can beprogrammed to any threshold voltage level within a threshold voltagewindow. The size of the threshold voltage window is delimited by theminimum and maximum threshold levels of the device, which in turncorrespond to the range of the charges that can be programmed onto thefloating gate. The threshold window generally depends on the memorydevice's characteristics, operating conditions and history. Eachdistinct, resolvable threshold voltage level range within the windowmay, in principle, be used to designate a definite memory state of thecell.

The transistor serving as a memory cell is typically programmed to a“programmed” state by one of two mechanisms. In “hot electroninjection,” a high voltage applied to the drain accelerates electronsacross the substrate channel region. At the same time a high voltageapplied to the control gate pulls the hot electrons through a thin gatedielectric onto the floating gate. In “tunneling injection,” a highvoltage is applied to the control gate relative to the substrate. Inthis way, electrons are pulled from the substrate to the interveningfloating gate.

The memory device may be erased by a number of mechanisms. For EPROM,the memory is bulk erasable by removing the charge from the floatinggate by ultraviolet radiation. For EEPROM, a memory cell is electricallyerasable, by applying a high voltage to the substrate relative to thecontrol gate so as to induce electrons in the floating gate to tunnelthrough a thin oxide to the substrate channel region (i.e.,Fowler-Nordheim tunneling.) Typically, the EEPROM is erasable byte bybyte. For flash EEPROM, the memory is electrically erasable either allat once or one or more blocks at a time, where a block may consist of512 bytes or more of memory.

Examples of Non-Volatile Memory Cells

The memory devices typically comprise one or more memory chips that maybe mounted on a card. Each memory chip comprises an array of memorycells supported by peripheral circuits such as decoders and erase, writeand read circuits. The more sophisticated memory devices also come witha controller that performs intelligent and higher level memoryoperations and interfacing. There are many commercially successfulnon-volatile solid-state memory devices being used today. These memorydevices may employ different types of memory cells, each type having oneor more charge storage element.

FIGS. 1A-1E illustrate schematically different examples of non-volatilememory cells.

FIG. 1A illustrates schematically a non-volatile memory in the form ofan EEPROM cell with a floating gate for storing charge. An electricallyerasable and programmable read-only memory (EEPROM) has a similarstructure to EPROM, but additionally provides a mechanism for loadingand removing charge electrically from its floating gate upon applicationof proper voltages without the need for exposure to UV radiation.Examples of such cells and methods of manufacturing them are given inU.S. Pat. No. 5,595,924.

FIG. 1B illustrates schematically a flash EEPROM cell having both aselect gate and a control or steering gate. The memory cell 10 has a“split-channel” 12 between source 14 and drain 16 diffusions. A cell isformed effectively with two transistors T1 and T2 in series. T1 servesas a memory transistor having a floating gate 20 and a control gate 30.The floating gate is capable of storing a selectable amount of charge.The amount of current that can flow through the T1's portion of thechannel depends on the voltage on the control gate 30 and the amount ofcharge residing on the intervening floating gate 20. T2 serves as aselect transistor having a select gate 40. When T2 is turned on by avoltage at the select gate 40, it allows the current in the T1's portionof the channel to pass between the source and drain. The selecttransistor provides a switch along the source-drain channel independentof the voltage at the control gate. One advantage is that it can be usedto turn off those cells that are still conducting at zero control gatevoltage due to their charge depletion (positive) at their floatinggates. The other advantage is that it allows source side injectionprogramming to be more easily implemented.

One simple embodiment of the split-channel memory cell is where theselect gate and the control gate are connected to the same word line asindicated schematically by a dotted line shown in FIG. 1B. This isaccomplished by having a charge storage element (floating gate)positioned over one portion of the channel and a control gate structure(which is part of a word line) positioned over the other channel portionas well as over the charge storage element. This effectively forms acell with two transistors in series, one (the memory transistor) with acombination of the amount of charge on the charge storage element andthe voltage on the word line controlling the amount of current that canflow through its portion of the channel, and the other (the selecttransistor) having the word line alone serving as its gate. Examples ofsuch cells, their uses in memory systems and methods of manufacturingthem are given in U.S. Pat. Nos. 5,070,032, 5,095,344, 5,315,541,5,343,063, and 5,661,053.

A more refined embodiment of the split-channel cell shown in FIG. 1B iswhen the select gate and the control gate are independent and notconnected by the dotted line between them. One implementation has thecontrol gates of one column in an array of cells connected to a control(or steering) line perpendicular to the word line. The effect is torelieve the word line from having to perform two functions at the sametime when reading or programming a selected cell. Those two functionsare (1) to serve as a gate of a select transistor, thus requiring aproper voltage to turn the select transistor on and off, and (2) todrive the voltage of the charge storage element to a desired levelthrough an electric field (capacitive) coupling between the word lineand the charge storage element. It is often difficult to perform both ofthese functions in an optimum manner with a single voltage. With theseparate control of the control gate and the select gate, the word lineneed only perform function (1), while the added control line performsfunction (2). This capability allows for design of higher performanceprogramming where the programming voltage is geared to the targeteddata. The use of independent control (or steering) gates in a flashEEPROM array is described, for example, in U.S. Pat. Nos. 5,313,421 and6,222,762.

FIG. 1C illustrates schematically another flash EEPROM cell having dualfloating gates and independent select and control gates. The memory cell10 is similar to that of FIG. 1B except it effectively has threetransistors in series. In this type of cell, two storage elements (i.e.,that of T1—left and T1—right) are included over its channel betweensource and drain diffusions with a select transistor T1 in between them.The memory transistors have floating gates 20 and 20′, and control gates30 and 30′, respectively. The select transistor T2 is controlled by aselect gate 40. At any one time, only one of the pair of memorytransistors is accessed for read or write. When the storage unit T1—leftis being accessed, both the T2 and T1—right are turned on to allow thecurrent in the T—left's portion of the channel to pass between thesource and the drain. Similarly, when the storage unit T1—right is beingaccessed, T2 and T1—left are turned on. Erase is effected by having aportion of the select gate polysilicon in close proximity to thefloating gate and applying a substantial positive voltage (e.g. 20V) tothe select gate so that the electrons stored within the floating gatecan tunnel to the select gate polysilicon.

FIG. 1D illustrates schematically a string of memory cells organizedinto an NAND chain. An NAND chain 50 consists of a series of memorytransistors M1, M2, . . . Mn (n=4, 8, 16 or higher) daisy-chained bytheir sources and drains. A pair of select transistors S1, S2 controlsthe memory transistors chain's connection to the external via the NANDchain's source terminal 54 and drain terminal 56. In a memory array,when the source select transistor S1 is turned on, the source terminalis coupled to a source line. Similarly, when the drain select transistorS2 is turned on, the drain terminal of the NAND chain is coupled to abit line of the memory array. Each memory transistor in the chain has acharge storage element to store a given amount of charge so as torepresent an intended memory state. A control gate of each memorytransistor provides control over read and write operations. A controlgate of each of the select transistors S1, S2 provides control access tothe NAND chain via its source terminal 54 and drain terminal 56respectively.

When an addressed memory transistor within an NAND chain is read andverified during programming, its control gate is supplied with anappropriate voltage. At the same time, the rest of the non-addressedmemory transistors in the NAND chain 50 are fully turned on byapplication of sufficient voltage on their control gates. In this way, aconductive path is effective created from the source of the individualmemory transistor to the source terminal 54 of the NAND chain andlikewise for the drain of the individual memory transistor to the drainterminal 56 of the chain. Memory devices with such NAND chain structuresare described in U.S. Pat. Nos. 5,570,315, 5,903,495, 6,046,935.

FIG. 1E illustrates schematically a non-volatile memory with adielectric layer for storing charge. Instead of the conductive floatinggate elements described earlier, a dielectric layer is used. Such memorydevices utilizing dielectric storage element have been described byEitan et al., “NROM: A Novel Localized Trapping, 2-Bit NonvolatileMemory Cell,” IEEE Electron Device Letters, vol. 21, no. 11, November2000, pp. 543-545. An ONO dielectric layer extends across the channelbetween source and drain diffusions. The charge for one data bit islocalized in the dielectric layer adjacent to the drain, and the chargefor the other data bit is localized in the dielectric layer adjacent tothe source. For example, U.S. Pat. Nos. 5,768,192 and 6,011,725 disclosea nonvolatile memory cell having a trapping dielectric sandwichedbetween two silicon dioxide layers. Multi-state data storage isimplemented by separately reading the binary states of the spatiallyseparated charge storage regions within the dielectric.

Memory Array

A memory device typically comprises of a two-dimensional array of memorycells arranged in rows and columns and addressable by word lines and bitlines. The array can be formed according to an NOR type or an NAND typearchitecture.

NOR Array

FIG. 2 illustrates an example of an NOR array of memory cells. Memorydevices with an NOR type architecture have been implemented with cellsof the type illustrated in FIG. 1B or 1C. Each row of memory cells areconnected by their sources and drains in a daisy-chain manner. Thisdesign is sometimes referred to as a virtual ground design. Each memorycell 10 has a source 14, a drain 16, a control gate 30 and a select gate40. The cells in a row have their select gates connected to word line42. The cells in a column have their sources and drains respectivelyconnected to selected bit lines 34 and 36. In some embodiments where thememory cells have their control gate and select gate controlledindependently, a steering line 30 also connects the control gates of thecells in a column.

Many flash EEPROM devices are implemented with memory cells where eachis formed with its control gate and select gate connected together. Inthis case, there is no need for steering lines and a word line simplyconnects all the control gates and select gates of cells along each row.Examples of these designs are disclosed in U.S. Pat. Nos. 5,172,338 and5,418,752. In these designs, the word line essentially performed twofunctions: row selection and supplying control gate voltage to all cellsin the row for reading or programming.

NAND Array

FIG. 3 illustrates an example of an NAND array of memory cells, such asthat shown in FIG. 1D. Along each column of NAND chains, a bit line iscoupled to the drain terminal 56 of each NAND chain. Along each row ofNAND chains, a source line may connect all their source terminals 54.Also the control gates of the NAND chains along a row are connected to aseries of corresponding word lines. An entire row of NAND chains can beaddressed by turning on the pair of select transistors (see FIG. 1D)with appropriate voltages on their control gates via the connected wordlines. When a memory transistor representing a memory cell within theNAND chain is being read, the remaining memory transistors in the chainare turned on hard via their associated word lines so that the currentflowing through the chain is essentially dependent upon the level ofcharge stored in the cell being read. An example of an NAND architecturearray and its operation as part of a memory system is found in U.S. Pat.Nos. 5,570,315, 5,774,397 and 6,046,935.

Block Erase

Programming of charge storage memory devices can only result in addingmore charge to its charge storage elements. Therefore, prior to aprogram operation, existing charge in a charge storage element must beremoved (or erased). Erase circuits (not shown) are provided to eraseone or more blocks of memory cells. A non-volatile memory such as EEPROMis referred to as a “Flash” EEPROM when an entire array of cells, orsignificant groups of cells of the array, is electrically erasedtogether (i.e., in a flash). Once erased, the group of cells can then bereprogrammed. The group of cells erasable together may consist of one ormore addressable erase unit. The erase unit or block typically storesone or more pages of data, the page being the unit of programming andreading, although more than one page may be programmed or read in asingle operation. Each page typically stores one or more sectors ofdata, the size of the sector being defined by the host system. Anexample is a sector of 512 bytes of user data, following a standardestablished with magnetic disk drives, plus some number of bytes ofoverhead information about the user data and/or the block in with it isstored.

Read/Write Circuits

In the usual two-state EEPROM cell, at least one current breakpointlevel is established so as to partition the conduction window into tworegions. When a cell is read by applying predetermined, fixed voltages,its source/drain current is resolved into a memory state by comparingwith the breakpoint level (or reference current I_(REF)). If the currentread is higher than that of the breakpoint level, the cell is determinedto be in one logical state (e.g., a “zero” state). On the other hand, ifthe current is less than that of the breakpoint level, the cell isdetermined to be in the other logical state (e.g., a “one” state). Thus,such a two-state cell stores one bit of digital information. A referencecurrent source, which may be externally programmable, is often providedas part of a memory system to generate the breakpoint level current.

In order to increase memory capacity, flash EEPROM devices are beingfabricated with higher and higher density as the state of thesemiconductor technology advances. Another method for increasing storagecapacity is to have each memory cell store more than two states.

For a multi-state or multi-level EEPROM memory cell, the conductionwindow is partitioned into more than two regions by more than onebreakpoint such that each cell is capable of storing more than one bitof data. The information that a given EEPROM array can store is thusincreased with the number of states that each cell can store. EEPROM orflash EEPROM with multi-state or multi-level memory cells have beendescribed in U.S. Pat. No. 5,172,338.

In practice, the memory state of a cell is usually read by sensing theconduction current across the source and drain electrodes of the cellwhen a reference voltage is applied to the control gate. Thus, for eachgiven charge on the floating gate of a cell, a corresponding conductioncurrent with respect to a fixed reference control gate voltage may bedetected. Similarly, the range of charge programmable onto the floatinggate defines a corresponding threshold voltage window or a correspondingconduction current window.

Alternatively, instead of detecting the conduction current among apartitioned current window, it is possible to set the threshold voltagefor a given memory state under test at the control gate and detect ifthe conduction current is lower or higher than a threshold current. Inone implementation the detection of the conduction current relative to athreshold current is accomplished by examining the rate the conductioncurrent is discharging through the capacitance of the bit line.

FIG. 4 illustrates the relation between the source-drain current I_(D)and the control gate voltage V_(CG) for four different charges Q1-Q4that the floating gate may be selectively storing at any one time. Thefour solid I_(D) versus V_(CG) curves represent four possible chargelevels that can be programmed on a floating gate of a memory cell,respectively corresponding to four possible memory states. As anexample, the threshold voltage window of a population of cells may rangefrom 0.5V to 3.5V. Six memory states may be demarcated by partitioningthe threshold window into five regions in interval of 0.5V each. Forexample, if a reference current, I_(REF) of 2 μA is used as shown, thenthe cell programmed with Q1 may be considered to be in a memory state“1” since its curve intersects with I_(REF) in the region of thethreshold window demarcated by V_(CG)=0.5V and 1.0V. Similarly, Q4 is ina memory state “5”.

As can be seen from the description above, the more states a memory cellis made to store, the more finely divided is its threshold window. Thiswill require higher precision in programming and reading operations inorder to be able to achieve the required resolution.

U.S. Pat. No. 4,357,685 discloses a method of programming a 2-stateEPROM in which when a cell is programmed to a given state, it is subjectto successive programming voltage pulses, each time adding incrementalcharge to the floating gate. In between pulses, the cell is read back orverified to determine its source-drain current relative to thebreakpoint level. Programming stops when the current state has beenverified to reach the desired state. The programming pulse train usedmay have increasing period or amplitude.

Prior art programming circuits simply apply programming pulses to stepthrough the threshold window from the erased or ground state until thetarget state is reached. Practically, to allow for adequate resolution,each partitioned or demarcated region would require at least about fiveprogramming steps to transverse. The performance is acceptable for2-state memory cells. However, for multi-state cells, the number ofsteps required increases with the number of partitions and therefore,the programming precision or resolution must be increased. For example,a 16-state cell may require on average at least 40 programming pulses toprogram to a target state.

FIG. 5 illustrates schematically a memory device with a typicalarrangement of a memory array 100 accessible by read/write circuits 170via row decoder 130 and column decoder 160. As described in connectionwith FIGS. 2 and 3, a memory transistor of a memory cell in the memoryarray 100 is addressable via a set of selected word line(s) and bitline(s). The row decoder 130 selects one or more word lines and thecolumn decoder 160 selects one or more bit lines in order to applyappropriate voltages to the respective gates of the addressed memorytransistor. Read/write circuits 170 are provided to read or write(program) the memory states of addressed memory transistors. Theread/write circuits 170 comprise a number of read/write modulesconnectable via bit lines to memory elements in the array.

Factors Affecting Read/Write Performance and Accuracy

In order to improve read and program performance, multiple chargestorage elements or memory transistors in an array are read orprogrammed in parallel. Thus, a logical “page” of memory elements areread or programmed together. In existing memory architectures, a rowtypically contains several interleaved pages. All memory elements of apage will be read or programmed together. The column decoder willselectively connect each one of the interleaved pages to a correspondingnumber of read/write modules. For example, in one implementation, thememory array is designed to have a page size of 532 bytes (512 bytesplus 20 bytes of overheads.) If each column contains a drain bit lineand there are two interleaved pages per row, this amounts to 8512columns with each page being associated with 4256 columns. There will be4256 sense modules connectable to read or write in parallel either allthe even bit lines or the odd bit lines. In this way, a page of 4256bits (i.e., 532 bytes) of data in parallel are read from or programmedinto the page of memory elements. The read/write modules forming theread/write circuits 170 can be arranged into various architectures.

As mentioned before, conventional memory devices improve read/writeoperations by operating in a massively parallel manner on all even orall odd bit lines at a time. This “alternate-bit-line” architecture of arow consisting of two interleaved pages will help to alleviate theproblem of fitting the block of read/write circuits. It is also dictatedby consideration of controlling bit-line to bit-line capacitivecoupling. A block decoder is used to multiplex the set of read/writemodules to either the even page or the odd page. In this way, wheneverone set bit lines are being read or programmed, the interleaving set canbe grounded to minimize immediate neighbor coupling.

However, the interleaving page architecture is disadvantageous in atleast three respects. First, it requires additional multiplexingcircuitry. Secondly, it is slow in performance. To finish read orprogram of memory cells connected by a word line or in a row, two reador two program operations are required. Thirdly, it is also not optimumin addressing other disturb effects such as field coupling betweenneighboring charge storage elements at the floating gate level when thetwo neighbors are programmed at different times, such as separately inodd and even pages.

United States Patent Publication No. 2004-0057318-A1 discloses a memorydevice and a method thereof that allow sensing a plurality of contiguousmemory cells in parallel. For example, all memory cells along a rowsharing the same word lines are read or programmed together as a page.This “all-bit-line” architecture doubles the performance of the“alternate-bit-line” architecture while minimizing errors caused byneighboring disturb effects. However, sensing all bit lines does bringup the problem of cross-talk between neighboring bit lines due inducedcurrents from their mutual capacitance. This is addressed by keeping thevoltage difference between each adjacent pair of bit lines substantiallyindependent of time while their conduction currents are being sensed.When this condition is imposed, all displacement currents due to thevarious bit lines' capacitance drop out since they all depend on a timevarying voltage difference. The sensing circuit coupled to each bit linehas a voltage clamp on the bit line so that the potential difference onany adjacent pair of connected bit lines is time-independent. With thebit line voltage clamped, the conventional method of sensing thedischarge due to the bit line capacitance can not be applied. Instead,the sensing circuit and method allow determination of a memory cell'sconduction current by noting the rate it discharges or charges a givencapacitor independent of the bit line. This will allow a sensing circuitindependent of the architecture of the memory array (i.e., independentof the bit line capacitance.) Especially, it allows the bit linevoltages to be clamped during sensing in order to avoid bit linecrosstalk.

As mentioned before, conventional memory devices improve read/writeoperations by operating in a massively parallel manner. This approachimproves performance but does have repercussions on the accuracy of readand write operations.

One issue is the source line bias error. This is particular acute formemory architecture where a large number memory cells have their sourcescoupled together in a source line to ground. Parallel sensing of thesememory cells with common source results in a substantial current throughthe source line. Owing to a non-zero resistance in the source line, thisin turn results in an appreciable potential difference between the trueground and the source electrode of each memory cell. During sensing, thethreshold voltage supplied to the control gate of each memory cell isrelative to its source electrode but the system power supply is relativeto the true ground. Thus sensing may become inaccurate due to theexistence of the source line bias error.

United States Patent Publication No. 2004-0057287-A1 discloses a memorydevice and a method thereof that allow sensing a plurality of contiguousmemory cells in parallel. The reduction in source line bias isaccomplished by read/write circuits with features and techniques formulti-pass sensing. When a page of memory cells are being sensed inparallel, each pass helps to identify and shut down the memory cellswith conduction current higher than a given demarcation current value.The identified memory cells are shut down by pulling their associatedbit lines to ground. In other words, those cells having higherconduction current and irrelevant to the present sensing are identifiedand have their current shut down before the actual data of the currentsensing is read.

Therefore there is a general need for high performance and high capacitynon-volatile memory with reduced power consumption. In particular, thereis a need for a compact non-volatile memory with enhanced read andprogram performance that is power efficient.

SUMMARY OF INVENTION

These needs for a high capacity and high performance non-volatile memorydevice are met by having a large page of read/write circuits to read andwrite a corresponding page of memory cells in parallel. In particular,interactive noises effects inherent in high density chip integration arethat may introduce errors into reading and programming are eithereliminated or minimized.

Source line bias is an error introduced by a non-zero resistance in theground loop of the read/write circuits. The error is caused by a voltagedrop across the resistance of the source path to the chip's ground whencurrent flows.

According to one aspect of the invention, when a page of memory cellsare sensed in parallel and their sources are coupled together to receivethe cell source signal at an aggregate access node, the operatingvoltage supplied to the bit line has the same reference point as theaggregate access node rather than the chip's ground. In this way anysource bias differences between the aggregate access node and the chip'sground will be tracked and compensated for in the supplied bit linevoltage.

According to another aspect of the invention, when a page of memorycells are sensed in parallel and their sources are coupled to the samepage source line, the operating voltage supplied to the bit line isreferenced with respect to an access node of the page source line ratherthan the chip's ground. In this way any source bias differences from theaccess node to the chip's ground will be tracked and compensated for inthe supplied bit line voltage.

According to yet another aspect of the invention, when a page of memorycells are sensed in parallel and their sources are coupled together toreceive the cell source signal at an aggregate access node, theoperating voltage supplied to the word line has the same reference pointas the aggregate access node rather than the chip's ground. In this wayany source bias differences between the aggregate access node and thechip's ground will be tracked and compensated for in the word linevoltage.

According to yet another aspect of the invention, when a page of memorycells are sensed in parallel and their sources are coupled to the samepage source line, the operating voltage supplied to the word line isreferenced with respect to an access node of the page source line ratherthan the chip's ground. In this way any source bias differences from theaccess node to the chip's ground will be tracked and compensated for inthe supplied word line voltage.

In one preferred voltage control circuit to track and compensate for thesource bias, the voltage control circuit references its base voltagewith respect to either the aggregate access node or the page accessnode. Its output voltage is generated by a reference current I_(REF)across an adjustable resistor. A cascode current mirror circuit 730 isemployed to maintain I_(REF) constant over the range of V_(BLC).

In another preferred voltage control circuit to track and compensate forthe source bias, the voltage control circuit references its base voltagewith respect to either the aggregate access node or the page accessnode. The control circuit uses a potential divider on a referencevoltage to obtain a desired output voltage. The reference voltage isdriven by a regulated output driver before having its output levelcontrolled by a DAC-controlled potential divider 840 to produce aprogrammed output voltage.

Additional features and advantages of the present invention will beunderstood from the following description of its preferred embodiments,which description should be taken in conjunction with the accompanyingdrawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A-1E illustrate schematically different examples of non-volatilememory cells.

FIG. 2 illustrates an example of an NOR array of memory cells.

FIG. 3 illustrates an example of an NAND array of memory cells, such asthat shown in FIG. 1D.

FIG. 4 illustrates the relation between the source-drain current and thecontrol gate voltage for four different charges Q1-Q4 that the floatinggate may be storing at any one time.

FIG. 5 illustrates schematically a typical arrangement of a memory arrayaccessible by read/write circuits via row and column decoders.

FIG. 6A illustrates schematically a compact memory device having a bankof read/write circuits, which provides the context in which the presentinvention is implemented.

FIG. 6B illustrates a preferred arrangement of the compact memory deviceshown in FIG. 6A.

FIG. 7A illustrates a conventional arrangement in which a bit linevoltage control, a word line voltage control and a source voltagecontrol are all referencing from the same ground of the IC memory chip.

FIG. 7B illustrates the error in both the gate voltage and drain voltageof a memory cell caused by a source line voltage drop.

FIG. 8 illustrates the effect of source bias errors in an examplepopulation distribution of a page of memory cells for a 4-state memory.

FIG. 9A illustrates an arrangement in which a bit line voltage controland/or a word line voltage control are compensated for source bias byhaving a reference point at the node where cell source signal accessesthe source lines, according to one preferred embodiment of theinvention.

FIG. 9B illustrates a bit line voltage control and a word line voltagecontrol are compensated for source bias by referencing with respect to apage source line, according to another preferred embodiment of theinvention.

FIG. 10 is a schematic diagram of a preferred sense module shown inFIGS. 9A and 9B that operates in combination with the tracking bit linevoltage control circuit to provide a bit line voltage compensated forsource bias.

FIG. 11 illustrates a preferred embodiment of the tracking bit linevoltage control circuit shown in FIGS. 9A and 9B.

FIG. 12 illustrates a preferred embodiment of the tracking word linevoltage control circuit shown in FIGS. 9A and 9B.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 6A illustrates schematically a compact memory device having a bankof read/write circuits, which provides the context in which the presentinvention is implemented. The memory device includes a two-dimensionalarray of memory cells 300, control circuitry 310, and read/writecircuits 370. The memory array 300 is addressable by word lines via arow decoder 330 and by bit lines via a column decoder 360. Theread/write circuits 370 is implemented as a bank of sense modules 480and allows a block (also referred to as a “page”) of memory cells to beread or programmed in parallel. In a preferred embodiment, a page isconstituted from a contiguous row of memory cells. In anotherembodiment, where a row of memory cells are partitioned into multipleblocks or pages, a block multiplexer 350 is provided to multiplex theread/write circuits 370 to the individual blocks.

The control circuitry 310 cooperates with the read/write circuits 370 toperform memory operations on the memory array 300. The control circuitry310 includes a state machine 312, an on-chip address decoder 314 and apower control module 316. The state machine 312 provides chip levelcontrol of memory operations. The on-chip address decoder 314 providesan address interface between that used by the host or a memorycontroller to the hardware address used by the decoders 330 and 370. Thepower control module 316 controls the power and voltages supplied to theword lines and bit lines during memory operations.

FIG. 6B illustrates a preferred arrangement of the compact memory deviceshown in FIG. 6A. Access to the memory array 300 by the variousperipheral circuits is implemented in a symmetric fashion, on oppositesides of the array so that access lines and circuitry on each side arereduced in half. Thus, the row decoder is split into row decoders 330Aand 330B and the column decoder into column decoders 360A and 360B. Inthe embodiment where a row of memory cells are partitioned into multipleblocks, the block multiplexer 350 is split into block multiplexers 350Aand 350B. Similarly, the read/write circuits are split into read/writecircuits 370A connecting to bit lines from the bottom and read/writecircuits 370B connecting to bit lines from the top of the array 300. Inthis way, the density of the read/write modules, and therefore that ofthe bank of sense modules 480, is essentially reduced by one half.

The entire bank of p sense modules 480 operating in parallel allows ablock (or page) of p cells along a row to be read or programmed inparallel. One example memory array may have p=512 bytes (512×8 bits). Inthe preferred embodiment, the block is a run of the entire row of cells.In another embodiment, the block is a subset of cells in the row. Forexample, the subset of cells could be one half of the entire row or onequarter of the entire row. The subset of cells could be a run ofcontiguous cells or one every other cell, or one every predeterminednumber of cells. Each sense module includes a sense amplifier forsensing the conduction current of a memory cell. A preferred senseamplifier is disclosed in United States Patent Publication No.2004-0109357-A1, the entire disclosure of which is hereby incorporatedherein by reference.

Source Line Error Management

One potential problem with sensing memory cells is source line bias.When a large number memory cells are sensed in parallel, their combinecurrents can result in significant voltage drop in a ground loop withfinite resistance. This results in a source line bias which causes errorin a sensing operation employing threshold voltage sensing. Also, if thecell is operating close to the linear region, the conduction current issensitive to the source-drain voltage once in that region, and thesource line bias will cause error in a sensing operation when the drainvoltage is offset by the bias.

FIG. 7A illustrates a conventional arrangement in which a bit linevoltage control, a word line voltage control and a source voltagecontrol are all referencing from the same ground of the IC memory chip.The read/write circuits 370 operate on a page of memory cellssimultaneously. Each sense module 480 in the read/write circuits iscoupled to a corresponding cell via a bit line, such as a bit line 36.For example, a sense module 480 senses the conduction current i₁(source-drain current) of a memory cell 10. The conduction current flowsfrom the sense module through the bit line 36 into the drain of thememory cell 10 and out from the source 14 before going through a sourceline 34 and a consolidated source line 40 and then to the chip's ground401 via a source control circuit 400. The source line 34 typically joinsall the sources of the memory cells in a page along a row in a memoryarray. In an integrated circuit chip, the source lines 34 of theindividual rows in a memory array are all tied together as multiplebranches of the consolidated source line 40 connected to the sourcecontrol circuit 400. The source control circuit 400 has a pull-downtransistor 402 controlled to pull the consolidated source line 40 to thechip's ground 401, which is ultimately connected to an external groundpad (e.g. Vss pad) of the memory chip. Even when metal strapping is usedto reduce the resistance of the source line, a non-zero resistance Rremains between the source electrode of a memory cell and the groundpad. Typically, the average ground loop resistance R can be as high as50 ohm.

For the entire page of memory being sensed in parallel, the totalcurrent flowing through the consolidated source line 40 is the sum ofall the conduction currents, i.e. I_(TOT)=i₁+i₂+ . . . , +i_(p).Generally each memory cell has a conduction current dependent on theamount of charge programmed into its charge storage element. For a givencontrol gate voltage of the memory cell, a smaller programmed chargewill yield a comparatively higher conduction current (see FIG. 4.) Whena finite resistance exists in the path between the source electrode of amemory cell and the ground pad, the voltage drop across the resistanceis given by V_(drop)˜i_(TOT)R.

For example, if 4,256 bit lines discharge at the same time, each with acurrent of 1 μA, then the source line voltage drop will be equal to 4000lines×1 μA/line×50 ohms ˜0.2 volts. This means instead of being atground potential, the effective source is now at 0.2V. Since the bitline voltage and the word line voltage are referenced with respect tothe same chip's ground 401, this source line bias of 0.2 volts will haveboth the effective drain voltage and control gate voltage reduced by0.2V.

FIG. 7B illustrates the error in the threshold voltage level of a memorycell caused by a source line voltage drop. The threshold voltage V_(T)supplied to the control gate 30 of the memory cell 10 is relative to thechip's ground 401. However, the effective V_(T) seen by the memory cellis the voltage difference between its control gate 30 and source 14.There is a difference of approximately V_(drop) or ΔV between thesupplied and effective V_(T) (ignoring the smaller contribution ofvoltage drop from the source 14 to the source line.) This ΔV or sourceline bias will contribute to a sensing error of, for example 0.2 voltswhen threshold voltages of the memory cells are sensed. This bias cannotbe easily removed as it is data-dependent, i.e., dependent on the memorystates of the memory cells of the page.

FIG. 7B also illustrates the error in the drain voltage level of amemory cell caused by a source line voltage drop. The drain voltageapplied to the drain 16 of the memory cell 10 is relative to the chip'sground 401. However, the effective drain voltage, V_(DS), seen by thememory cell is the voltage difference between its drain 16 and source14. There is a difference of approximately ΔV between the supplied andeffective V_(DS). This ΔV or source line bias will contribute to asensing error when the memory cells are sensed in an operating regionsensitive to V_(DS). As described above, this bias cannot be easilyremoved as it is data-dependent, i.e., dependent on the memory states ofthe memory cells of the page.

FIG. 8 illustrates the effect of source bias errors in an examplepopulation distribution of a page of memory cells for a 4-state memory.Each cluster of memory state is programmed within a range of conductioncurrents I_(SD) clearly separated from each other. For example, abreakpoint 381 is a demarcating current value between two clusters,respectively representing the “1” and “2” memory states. A necessarycondition for a “2” memory state will be that it has a conductioncurrent less than the breakpoint 381. If there were no source line bias,the population distribution with respect to the supplied thresholdvoltage V_(T) will be given by the curve with the solid line. However,because of the source line bias error, the effective threshold voltageof each of the memory cells at its control gate is reduced from thesupplied voltage relative to ground by the source line bias ΔV.Similarly, the effective drain voltage is also reduced from the suppliedvoltage by the source line bias.

The source line bias results in a shifting of the distribution (brokenline) towards a higher supplied V_(T) to make up for the shortfall inthe effective voltage. The shifting will be more for that of the higher(lower current) memory states. If the breakpoint 381 is designed for thecase without source line error, then the existence of a source lineerror will have some of the tail end of “ 1” states having conductioncurrents to appear in a region of no conduction, which means higher thanthe breakpoint 381. This will result in some of the “1” states (moreconducting) being mistakenly demarcated as “2” states (less conducting.)

Drain Compensation of Source Line Bias

According to one aspect of the invention, when a page of memory cellsare sensed in parallel and their sources are coupled together to receivethe cell source signal at an aggregate access node, the operatingvoltage supplied to the bit line has the same reference point as theaggregate access node rather than the chip's ground. In this way anysource bias differences between the aggregate access node and the chip'sground will be tracked and compensated for in the supplied bit linevoltage.

Generally, the source path from each memory cell to the chip's groundvaries over a range since each memory cell will have a different networkpath to the chip's ground. Also the conduction current of each memorycell depends on the data programmed into it. Even among the memory cellsof a page, there will be some variations in the source bias. However,when the reference point is taken as close to the memory cells' sourcesas possible, the errors will at least be minimized.

FIG. 9A illustrates an arrangement in which a bit line voltage controland/or a word line voltage control are compensated for source bias byhaving a reference point at the node where cell source signal accessesthe source lines, according to one preferred embodiment of theinvention. Similar to FIG. 7A, the read/write circuits 370 operate on apage of memory cells simultaneously. Each sense module 480 in theread/write circuits is coupled to a corresponding cell via a bit line,such as a bit line 36. A page source line 34 is coupled to the source ofeach memory cell of the page along a row in the memory array. Multiplerows have their page source lines coupled together and to the sourcecontrol circuit 400 via an aggregate access node 35. The source controlcircuit 400 has a pull-down transistor 402 controlled to pull theaggregate access node 35 and therefore the page source line 34 to thechip's ground 401 through a ground path formed by a consolidated sourceline with resistance R_(S). The ground 401 is ultimately connected to anexternal ground pad (e.g. Vss pad) of the memory chip. Thus, the sourcecontrol circuit 400 controls the cell source signal at the aggregateaccess node 35. Due to the finite resistance ground path, the cellsource signal is not at 0V but has a source bias of ΔV₁.

A bit line voltage control embodied as a tracking bit line voltage clamp700 is implemented to compensate for the data dependent source bias.This is accomplished by generating an output voltage V_(BLC) in anoutput 703 that is referencing at the same point as the cell sourcesignal at the aggregate access node 35 instead of the external groundpad. In this way, at least the source bias due to the resistance R_(S)of the consolidated source line is eliminated.

According to another aspect of the invention, when a page of memorycells are sensed in parallel and their sources are coupled to the samepage source line, the operating voltage supplied to the bit line isreferenced with respect to an access node of the page source line ratherthan the chip's ground. In this way any source bias differences from thepage access node to the chip's ground will be tracked and compensatedfor in the supplied bit line voltage.

FIG. 9B illustrates a bit line voltage control and a word line voltagecontrol are compensated for source bias by referencing with respect to apage source line, according to another preferred embodiment of theinvention.

The arrangement is similar to that of FIG. 9A except the reference pointfor the bit line voltage control 700 and word line voltage control 800is now taken essentially at the selected page source line. A page sourceline multiplexor 780 is used to selectively couple the selected pagesource line to a page access node 37, which serves as the referencepoint.

A bit line voltage control embodied as a tracking bit line voltage clamp700 is implemented to compensate for the data dependent source bias.This is accomplished by generating an output voltage V_(BLC) in anoutput 703 that is referencing with respect to the voltage at the accessnode 38 of the page source line 34 instead of referencing to theexternal ground pad. In this way, the source bias is better correcteddue the location of the reference point at the access node 37, which isspecific to the page.

FIG. 10 is a schematic diagram of a preferred sense module shown inFIGS. 9A and 9B that operates in combination with the tracking bit linevoltage control circuit to provide a bit line voltage compensated forsource bias. In the example shown, the sense module 480 senses theconduction current of a memory cell in a NAND chain 50 via a coupled bitline 36. It has a sense node 481 that can be selectively coupled to abit line, a sense amplifier 600 or a readout bus 499. Initially, anisolation transistor 482, when enabled by a signal BLS connects the bitline 36 to the sense node 481. The sense amplifier 600 senses the sensenode 481. The sense amplifier includes a precharge/clamp circuit 640, acell current discriminator 650 and a latch 660.

The sense module 480 enables the conduction current of the selectedmemory cell in the NAND chain to be sensed. The conduction current is afunction of the charge programmed into the memory cell and the appliedV_(T)(i) when there exists a nominal voltage difference between thesource and drain of the memory cell. Prior to sensing, the voltages tothe gates of the selected memory cell must be set via the appropriateword lines and bit line.

The precharge operation starts with the unselected word line charging toa voltage Vread followed by charging the selected word line to apredetermined threshold voltage V_(T)(i) for a given memory state underconsideration.

Then the precharged circuit 640 brings the bit line voltage to apredetermined drain voltage appropriate for sensing. This will induce asource-drain conduction current to flow in the selected memory cell inthe NAND chain 50, which is detected from the channel of the NAND chainvia a coupled bit line 36.

When the V_(T)(i) voltage is stable, the conduction current or theprogrammed threshold voltage of the selected memory cell can be sensedvia the coupled bit line 36. The sense amplifier 600 is then coupled tothe sense node to sense the conduction current in the memory cell. Thecell current discriminator 650 serves as a discriminator or comparatorof current levels. It effectively determines whether the conductioncurrent is higher or lower than a given demarcation current value I₀(j).If it is higher, the latch 660 is set to a predetermined state with thesignal INV=1.

A pull-down circuit 486 is activated in response to the latch 660setting the signal INV to HIGH. This will pull down the sense node 481and therefore the connected bit line 36 to ground voltage. This willinhibit the conduction current flow in the memory cell 10 irrespectiveof the control gate voltage since there will be no voltage differencebetween its source and drain.

As shown in FIGS. 9A and 9B, there will be a page of memory cells beingoperated on by a corresponding number of sense modules 480. A pagecontroller 498 supplies control and timing signals to each of the sensemodules. The page controller 498 cycles each of the sense module 480through a predetermined sequence of operations and also supplies apredetermined demarcation current value I₀(j) during the operations. Asis well known in the arts, the demarcation current value can also beimplemented as a demarcation threshold voltage, or time period forsensing. After the last pass, the page controller 498 enables a transfergate 488 with a signal NCO to read the state of the sense node 481 assensed data to a readout bus 499. In all, a page of sense data will beread out from all the multi-pass modules 480. Similar sense modules havebeen disclosed in U.S. patent application Ser. No. 11/015,199 filed Dec.16, 2004 by Cernea et al., entitled “IMPROVED MEMORY SENSING CIRCUIT ANDMETHOD FOR LOW VOLTAGE OPERATION”. The entire disclosure of U.S. patentapplication Ser. No. 11/015,199 is herein incorporated by reference.

The sense module 480 incorporates a constant voltage supply andmaintains the bit line at constant voltage during sensing in order toavoid bit line to bit line coupling. This is preferably implemented bythe bit line voltage clamp 610. The bit line voltage clamp 610 operateslike a diode clamp with a transistor 612 in series with the bit line 36.Its gate is biased to a constant voltage V_(BLC) equal to the desiredbit line voltage V_(BL) above its threshold voltage V_(TN). In this way,it isolates the bit line from the sense node 481 and set a constantvoltage level for the bit line, such as the desired V_(BL)=0.4 to 0.7volts. In general the bit line voltage level is set to a level such thatit is sufficiently low to avoid a long precharge time, yet sufficientlyhigh to avoid ground noise and other factors such as operating in thesaturated region where V_(DC) is above 0.2 volts.

Thus, when operating at a low V_(BL), especially one that approachingthe linear region, it is important that V_(BL) is accurately rendered,as small variations can lead to significant changes in conductioncurrents. This means V_(BLC)=V_(BL)+V_(TN) must be accurately set tominimize the source line bias.

FIG. 11 illustrates a preferred embodiment of the tracking bit linevoltage control circuit shown in FIGS. 9A and 9B. The tracking bit linevoltage control circuit 700 basically provides an output voltage V_(BLC)on an output line 703. The output voltage is essentially generated by areference current I_(REF) across an adjustable resistor R 720. A cascodecurrent mirror circuit 730 is employed to maintain I_(REF) constant overthe range of V_(BLC). The cascode current mirror circuit 730 has twobranches, with a first branch formed by two n-transistors 732, 734connected as diodes in series and a second, mirrored branch formed bytwo other n-transistors 736, 738 connected in series. The gates of thetransistors 732 and 736 are interconnected, and the gates of thetransistors 734 and 738 are interconnected. An I_(REF) source isconnected to the drain of the transistor 732 so that I_(REF) flows downthe first branch and is also mirrored in the second branch. A V_(HIGH)source is connected to the drain of the transistor 736. The sources ofthe transistors 734 and 738 are interconnected to form a base rail 701.

The output voltage is taken from a tap between the serially connectedtransistors 736 and 738. If the voltage of the base rail 701 is at V1,then V_(BLC)=V1+V_(TN). This is because the voltage on the drain of thetransistor 734 is V1 plus a threshold voltage of the n-transistor, andthe same I_(REF) is also mirrored in the second branch, resulting in thesame voltage appearing on the drain of the transistor 738.

The voltage V1 at the base rail 701 is set by the voltage drop acrossthe resistor R 720 due to the current 2I_(REF) plus a base voltage atthe node 721. The base voltage at node the 721 is selectable by a basevoltage selector 740. The base voltage selector 740 selectively connectsthe node 721 to the aggregate access node 35 (see FIG. 9A) or to thepage access node 37 of the page source line (see FIG. 9B) via atransistor 742 when a control signal ConSL is asserted at its gate.Alternatively, the selector circuit 720 selectively connects the node721 to ground 401 via a transistor 744 when a control signal ConGND isasserted at its gate. Thus, it will be seen that when the signal ConSLis asserted, V1=ΔV₁+2I_(REF)R, and the output of the tracking bit linevoltage control circuit, V_(BLC)=ΔV₁+2I_(REF)R+V_(TN). In the case ofcontrolling the bit line voltage clamp 610 (see FIG. 10), then-transistor 734 is chosen to have the same V_(TN) as that of thetransistor forming the bit line voltage clamp 610. The resistor R isthen adjusted so that the desired bit line voltage V_(BL) is set by2I_(REF)R. By referencing with respect to the aggregate access node 35or the page access node 37, a significant portion of source bias ΔV₁that is above ground potential will be compensated automatically inV_(BLC).

Control Gate Compensation of Source Line Bias

According to yet another aspect of the invention, when a page of memorycells are sensed in parallel and their sources are coupled together toreceive the cell source signal at an aggregate access node, theoperating voltage supplied to the word line has the same reference pointas the aggregate access node rather than the chip's ground. In this wayany source bias differences between the aggregate access node and thechip's ground will be tracked and compensated for in the supplied wordline voltage.

As shown in FIG. 9A, a word line voltage control embodied as a trackingword line voltage clamp 800 is implemented to compensate for the datadependent source bias. This is accomplished by generating an outputvoltage V_(WL) in an output 803 that is referencing at the same point asthe cell source signal at the aggregate node 35 instead of the externalground pad. In this way, at least the source bias due to the resistanceof the consolidated source line (see FIG. 7A) is eliminated.

According to yet another aspect of the invention, when a page of memorycells are sensed in parallel and their sources are coupled to the samepage source line, the operating voltage supplied to the word line isreferenced with respect to an access node of the page source line ratherthan the chip's ground. In this way any source bias differences from thepage access node to the chip's ground will be tracked and compensatedfor in the supplied word line voltage.

As shown in FIG. 9B, a word line voltage control embodied as a trackingword line voltage clamp 800 is implemented to compensate for the datadependent source bias. This is accomplished by generating an outputvoltage V_(WL) in an output 803 that is referencing at the same point asthe access node 38 to the selected page source line instead of theexternal ground pad. In this way, the source bias is better correcteddue the location of the reference point at the access node 38, which isspecific to the page.

FIG. 12 illustrates a preferred embodiment of the tracking word linevoltage control circuit shown in FIGS. 9A and 9B. The tracking word linevoltage control circuit 800 essentially uses a potential divider on areference voltage to obtain a desired output voltage V_(WL) on an output803. A reference voltage V_(REF) is provided by a VREF circuit 820.V_(REF) is driven by a regulated output driver 830. The output level ofthe driven V_(REF) is controlled by a DAC-controlled potential divider840 to produce a programmed V_(WL) at the output 803.

The regulated output driver 830 includes a p-transistor 832 driving anoutput from a comparator 834. The drain of the p-transistor 832 isconnected to a voltage source, V_(HIGH) and its gate is controlled bythe output of the comparator 834. The comparator 834 receives VREF atits “−” terminal and compares it with a signal fed back from the sourceof the p-transistor. Also, a capacitor 836 is used to AC couple theoutput of the comparator with the “+” terminal. If the voltage at thesource of the p-transistor 832 is less than V_(REF), the output of thecomparator is low, turning on the p-transistor 832, which results in thevoltage at the source rising to the level of V_(REF). On the other hand,if V_(REF) is exceeded, the comparator output will turn off thep-transistor 832 to effect regulation, so that a driven, regulatedV_(REF) appears across the potential divider 840. The potential divider840 is formed by a series of resistors; each tap between any tworesistors is switchable to the output 803 by a transistor such astransistor 844 that is turned on by a signal such as DAC1. In this way,by selectively connecting the output 803 to a tap in the potentialdivider, a desired fraction of V_(REF) can be obtained; i.e.,(n*r/r_(TOT))*V_(REF), where n is the number of r DAC setting selected.

V_(REF) and therefore V_(WL) are referenced with respect to a node 821.The base voltage at the node 821 is selectable by a base voltageselector 850. The base voltage selector 740 selectively connects thenode 721 to the aggregate access node 35 (see FIG. 9A) or to the pageaccess node 37 of the page source line (see FIG. 9B) via a transistor742 when a control signal ConSL is asserted at its gate. Alternatively,the selector circuit 850 selectively connects the node 821 to ground 401via a transistor 854 when a control signal ConGND is asserted at itsgate. Thus, it will be seen that when the signal ConSL is asserted, ΔV₁will appear at the node 821, which will become the base voltage for theVREF circuit 820 and the voltage divider 840. Therefore the output ofthe tracking word line voltage control circuit 800 will haveV_(WL)=(n*r/r_(TOT))*V_(REF)+ΔV₁. By referencing with respect to theaggregate access node 35 or the page access node 37, a significantportion of source bias ΔV₁ that is above ground potential will becompensated automatically in V_(WL).

The tracking voltage control circuit 800 can alternatively be employedto track the source bias for the V_(BLC) used in controlling the bitline voltage clamp 610 (see FIG. 10). Essentially, the output voltage isset to provide V_(BL)+V_(TN)+ΔV₁.

Although the various aspects of the present invention have beendescribed with respect to certain embodiments, it is understood that theinvention is entitled to protection within the full scope of theappended claims.

1. In a non-volatile memory device having individual pages of memorycells to be sensed in parallel, each memory cell having a source, adrain, a charge storage unit and a control gate for controlling aconduction current along said drain and source, a method of sensing apage of memory cells, comprising: providing a page source line to eachindividual page; coupling the source of each memory cell of eachindividual page to its page source line; coupling the page source linesof individual pages to an aggregate node for connection to a sourcevoltage control circuit for sensing operation; coupling the drain ofeach memory cell of said page to an associated bit line; and providing apredetermined bit line voltage to the associated bit line of each memorycell of said page for sensing operation, wherein each said predeterminedbit line voltage is referenced with respect to said aggregate node so asnot to be affected by any voltage differences between said aggregatenode and a ground reference.
 2. The method of sensing as in claim 1,wherein said page source line is at a higher potential than that of saidsource voltage control circuit.
 3. The method of sensing as in claim 1,wherein said source voltage control circuit is referenced with respectto said ground reference.
 4. The method of sensing as in claim 1,wherein said providing a predetermined bit line voltage furthercomprises: providing a bit line voltage clamp; and generating a controlvoltage to control the bit line voltage clamp to clamp the bit line atthe predetermined bit line voltage.
 5. The method of sensing as in claim4, wherein said generating a control voltage further comprises:providing a reference current; providing a predetermined resistor; andgenerating the control voltage by passing the reference current acrossthe predetermined resistor.
 6. The method of sensing as in claim 4,wherein said providing a predetermined bit line voltage clamp includes:providing a first transistor having a source, drain and control gate andhaving a diode voltage drop across the control gate and the source;coupling a voltage supply to the drain of the first transistor; couplingan associated bit line to the source of the first transistor; andapplying the control voltage to the control gate of the first transistorsuch that the control voltage has a value given by the predetermined bitline voltage plus the diode voltage drop.
 7. The method of sensing as inclaim 6, wherein said generating a control voltage further comprises:providing a reference current; providing a predetermined resistor;providing a second transistor having a substantially similar diodevoltage drop as that of the first transistor; and generating the controlvoltage by summing the voltage drop across the second transistor and thevoltage drop created by passing the reference current across thepredetermined resistor.
 8. The method of sensing as in claim 6, whereinsaid generating a control voltage further comprises: providing aregulated reference voltage; providing a DAC-controlled potentialdivider; and generating the control voltage by dividing said regulatedreference voltage using the DAC-controlled potential divider.
 9. Themethod as in any one of claims 1-8, wherein each of said memory cellsstores one bit of data.
 10. The method as in any one of claims 1-8,wherein each of said memory cells stores more than one bit of data. 11.In a non-volatile memory device having individual pages of memory cellsto be sensed in parallel, each memory cell having a source, a drain, acharge storage unit and a control gate for controlling a conductioncurrent along said drain and source, said memory device, comprising: apage source line coupled to the source of each memory cell in a page; anaggregate node coupled to individual page source lines; a source voltagecontrol circuit coupled via said aggregate node to a page source line ofa selected page for memory operation; an associated bit line coupling tothe drain of each memory cell of said page; and a bit line voltagesupply for providing a predetermined bit line voltage to the associatedbit line of each memory cell of said page for sensing operation, whereineach said predetermined bit line voltage is referenced with respect tosaid aggregate node so as not to be affected by any voltage differencesbetween said aggregate node and a ground reference.
 12. The memorydevice as in claim 11, wherein said page source line is at a higherpotential than that of said source voltage control circuit.
 13. Thememory device as in claim 11, wherein said source voltage controlcircuit is referenced with respect to said ground reference.
 14. Thememory device as in claim 11, wherein said bit line voltage supplyfurther comprises: a bit line voltage clamp; and a control voltagegenerator for generating a control voltage to control the bit linevoltage clamp to clamp the bit line at the predetermined bit linevoltage.
 15. The memory device as in claim 14, wherein said controlvoltage generator further comprises: a reference current; apredetermined resistor; and the control voltage generated is a functionof the reference current and the predetermined resistor.
 16. The memorydevice as in claim 14, wherein said bit line voltage clamp includes: afirst transistor having a source, drain and control gate and having adiode voltage drop across the control gate and the source; a voltagesupply coupled to the drain of the first transistor; an associated bitline coupled to the source of the first transistor; and the controlvoltage applied to its control gate such that the control voltage have avalue given by the predetermined bit line voltage plus the diode voltagedrop.
 17. The memory device as in claim 16, wherein said control voltagegenerator further comprises: a reference current; a predeterminedresistor; a second transistor having a substantially similar diodevoltage drop as that of the first transistor; and the control voltagegenerated is a function of the voltage drop across the second transistorand the voltage drop created by passing the reference current across thepredetermined resistor.
 18. The memory device as in claim 16, whereinsaid control voltage generator further comprises: a regulated referencevoltage; a DAC-controlled potential divider; and an output controlvoltage given by dividing said regulated reference voltage using theDAC-controlled potential divider.
 19. The memory device as in any one ofclaims 11-18, wherein each of said memory cells stores one bit of data.20. The memory device as in any one of claims 11-18, wherein each ofsaid memory cells stores more than one bit of data.
 21. In anon-volatile memory device having individual pages of memory cells to besensed in parallel, each memory cell having a source, a drain, a chargestorage unit and a control gate for controlling a conduction currentalong said drain and source, a method of sensing a page of memory cells,comprising: providing a page source line; coupling the source of eachmemory cell of said page to said page source line; switching said pagesource line to a source voltage control circuit for sensing operation;coupling the drain of each memory cell of said page to an associated bitline; and providing a predetermined bit line voltage to the associatedbit line of each memory cell of said page for sensing operation, whereineach said predetermined bit line voltage is referenced with respect tosaid page source line of said page so as not to be affected by anyvoltage differences between the page source line and a ground reference.22. The method of sensing as in claim 21, wherein said page source lineis at a higher potential than that of said source voltage controlcircuit.
 23. The method of sensing as in claim 21, wherein said sourcevoltage control circuit is referenced with respect to said groundreference.
 24. The method of sensing as in claim 21, wherein saidproviding a predetermined bit line voltage further comprises: providinga bit line voltage clamp; and generating a control voltage to controlthe bit line voltage clamp to clamp the bit line at the predeterminedbit line voltage.
 25. The method of sensing as in claim 24, wherein saidgenerating a control voltage further comprises: providing a referencecurrent; providing a predetermined resistor; and generating the controlvoltage by passing the reference current across the predeterminedresistor.
 26. The method of sensing as in claim 24, wherein saidproviding a predetermined bit line voltage clamp includes: providing afirst transistor having a source, drain and control gate and having adiode voltage drop across the control gate and the source; coupling avoltage supply to the drain of the first transistor; coupling the bitline to its source; and applying the control voltage to its control gatesuch that the control voltage have a value given by the predeterminedbit line voltage plus the diode voltage drop.
 27. The method of sensingas in claim 26, wherein said generating a control voltage furthercomprises: providing a reference current; providing a predeterminedresistor; providing a second transistor having a substantially similardiode voltage drop as that of the first transistor; and generating thecontrol voltage by summing the voltage drop across the second transistorand the voltage drop created by passing the reference current across thepredetermined resistor.
 28. The method of sensing as in claim 26,wherein said generating a control voltage further comprises: providing aregulated reference voltage; providing a DAC-controlled potentialdivider; and generating the control voltage by dividing said regulatedreference voltage using the DAC-controlled potential divider.
 29. Themethod as in any one of claims 21-28, wherein each of said memory cellsstores one bit of data.
 30. The method as in any one of claims 21-28,wherein each of said memory cells stores more than one bit of data. 31.In a non-volatile memory device having individual pages of memory cellsto be sensed in parallel, each memory cell having a source, a drain, acharge storage unit and a control gate for controlling a conductioncurrent along said drain and source, said memory device, comprising: apage source line coupled to the source of each memory cell in a page; apage source line multiplexor; a source voltage control circuit coupledvia said page source line multiplexor to a page source line of aselected page for memory operation; an associated bit line coupling tothe drain of each memory cell of said page; and a bit line voltagesupply for providing a predetermined bit line voltage to the associatedbit line of each memory cell of said page for sensing operation, whereineach said predetermined bit line voltage is referenced with respect tosaid page source line so as not to be affected by any voltagedifferences between said page source line and a ground reference. 32.The memory device as in claim 31, wherein said page source line is at ahigher potential than that of said source voltage control circuit. 33.The memory device as in claim 31, wherein said source voltage controlcircuit is referenced with respect to said ground reference.
 34. Thememory device as in claim 31, wherein said bit line voltage supplyfurther comprises: a bit line voltage clamp; and a control voltagegenerator for generating a control voltage to control the bit linevoltage clamp to clamp the bit line at the predetermined bit linevoltage.
 35. The memory device as in claim 34, wherein said controlvoltage generator further comprises: a reference current; apredetermined resistor; and the control voltage generated is a functionof the reference current and the predetermined resistor.
 36. The memorydevice as in claim 34, wherein said bit line voltage clamp includes: afirst transistor having a source, drain and control gate and having adiode voltage drop across the control gate and the source; a voltagesupply coupled to the drain of the first transistor; an associated bitline coupled to the source of the first transistor; and the controlvoltage applied to its control gate such that the control voltage have avalue given by the predetermined bit line voltage plus the diode voltagedrop.
 37. The memory device as in claim 36, wherein said control voltagegenerator further comprises: a reference current; a predeterminedresistor; a second transistor having a substantially similar diodevoltage drop as that of the first transistor; and the control voltagegenerated is a function of the voltage drop across the second transistorand the voltage drop created by passing the reference current across thepredetermined resistor.
 38. The memory device as in claim 36, whereinsaid control voltage generator further comprises: a regulated referencevoltage; a DAC-controlled potential divider; and an output controlvoltage given by dividing said regulated reference voltage using theDAC-controlled potential divider.
 39. The memory device as in any one ofclaims 31-38, wherein each of said memory cells stores one bit of data.40. The memory device as in any one of claims 31-38, wherein each ofsaid memory cells stores more than one bit of data.